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rev. 1.3 december 2007 www.aosmd.com page 1 of 14 AOZ1210 ezbuck 2a simple buck regulator general description the AOZ1210 is a high ef?iency, simple to use, 2a buck regulator ?xible enough to be optimized for a variety of applications. the AOZ1210 works from a 4.5v to 27v input voltage range, and provides up to 2a of continuous output current on each buck regulator output. the output v oltage is adjustable down to 0.8v. features 4.5v to 27v operating input voltage range 70m ? internal nfet, ef?iency: up to 95% internal soft start output voltage adjustable down to 0.8v 2a continuous output current fixed 370khz pwm operation cycle-by-cycle current limit short-circuit protection thermal shutdown small size so-8 packages applications p oint of load dc/dc conversion set top boxes d vd drives and hdd lcd monitors & tvs cable modems t elecom/networking/datacom equipment t ypical application figure 1. 3.3v/2a buck regulator lx vin bs vin vout fb gnd en vbias comp c1 22f c7 c4 c4, c6 22f r1 r2 r c c c l1 6.8h AOZ1210
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 2 of 14 ordering information all aos products are offering in packaging with pb-free plating and compliant to rohs standards. please visit www.aosmd.com/web/quality/rohs_compliant.jsp for additional information. pin con?uration pin description pa rt number ambient temperature range package environmental a oz1210ai -40? to +85? so-8 rohs vbias vin en comp 1 2 3 4 lx bst gnd fb so-8 (top view) 8 7 6 5 pin number pin name pin function 1l x pwm output connection to inductor. lx pin needs to be connected externally. thermal connection f or output stage. 2 bst bootstrap voltage input. high side driver supply. connected to 0.1? capacitor between bst and lx. 3 gnd ground. 4f bf eedback input. it is regulated to 0.8v. the fb pin is used to determine the pwm output voltage via a resistor divider between the output and gnd. 5 comp external loop compensation. output of internal error ampli?r. connect a series rc network to gnd for control loop compensation. 6e n enable pin. the enable pin is active high. connect en pin to v in if not used. do not leave the en pin ?ating. 7v in supply voltage input. range from 4.5v to 27v. when v in rises above the uvlo threshold the device starts up. all v in pins need to be connected externally. 8 vbias compensation pin of internal linear regulator. place put a 1? capacitor between this pin and g round.
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 3 of 14 block diagram absolute maximum ratings exceeding the absolute maximum ratings may damage the device. note: 1. devices are inherently esd sensitive, handling precautions are required. human body model rating: 1.5k ? in series with 100pf. recommend operating ratings the device is not guaranteed to operate beyond the maximum operating ratings. note: 2. the value of ja is measured with the device mounted on 1-in 2 fr-4 board with 2oz. copper, in a still air environment with t a = 25?. the value in any given application depends on the user's speci? board design. 370khz/24khz oscillator gnd vin en vbias fb comp lx bst otp ilimit pwm control logic 5v ldo regulator uvlo & por +5v gm = 200 a/v softstart reference & bias 0.8v q1 q2 pwm comp isen eamp 0.2v + + + + + frequency foldback comparator p arameter rating supply voltage (v in ) 30v lx to gnd -0.7v to v in +0.3v en to gnd -0.3v to v in +0.3v fb to gnd -0.3v to 6v comp to gnd -0.3v to 6v bst to gnd v lx +6v vbias to gnd -0.3v to 6v j unction temperature (t j ) +150? storage temperature (t s ) -65? to +150? esd rating: human body model (1) 2kv p arameter rating supply voltage (v in ) 4.5v to 27v output voltage range 0.8v to v in ambient temperature (t a ) -40? to +85? pa c kage thermal resistance so-8 ( ja ) (2 ) 105?/w
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 4 of 14 electrical characteristics t a = 25?, v in = v en = 12v, v out = 3.3v unless otherwise speci?d (3 ) note: 3. speci?ation in bold indicate an ambient temperature range of -40? to +85?. these speci?ations are guaranteed by design. symbol p arameter conditions min. t yp. max. units v in supply voltage 4.5 27 v v uvlo input under-voltage lockout threshold v in rising v in falling 4.3 4.1 v i in supply current (quiescent) i out = 0, v fb = 1.2v, v en > 2v 23 ma i off shutdown supply current v en = 0v 320 ? v fb f eedback voltage 0.782 0.8 0.818 v load regulation 0.5 % line regulation 0.08 % / v i fb f eedback voltage input current 200 na enable v en en input threshold off threshold on threshold 2.5 0.6 v v hys en input hysteresis 200 mv modulator f o f requency 315 370 425 khz d max maximum duty cycle 85 % d min minimum duty cycle 6% g vea error ampli?r voltage gain 500 v / v g ea error ampli?r transconductance 200 ? / v protection i lim current limit 3.5 5.0 a over-temperature shutdown limit t j rising t j falling 145 100 ? f sc short circuit hiccup frequency v fb = 0v 24 khz t ss soft start interval 4ms pwm output stage r ds(on) high-side switch on-resistance 70 100 m ? high-side switch leakage v en = 0v, v lx = 0v 10 ?
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 5 of 14 t ypical performance characteristics circuit of figure 1. t a = 25?, v in = v en = 24v, v out = 3.3v unless otherwise speci?d. light load (dcm) operation full load (ccm) operation startup to full load short circuit protection 50% to 100% load transient short circuit recovery 1 s/div 1 s/div 2ms/div 200 s/div 200 s/div 2ms/div vin ripple 0.1v/div vo ripple 20mv/div vo 2v/div lin 0.5a/div vo ripple 200mv/div lo 1a/div vo 2v/div ll 2a/div vo 2v/div il 2a/div il 1a/div vlx 20v/div vin ripple 0.1v/div vo ripple 20mv/div il 1a/div vlx 20v/div
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 6 of 14 ef?iency curves efficiency v in = 5v 75 80 85 90 95 1.8v output 5.0v output 5.0v output 3.3v output 3.3v output 8.0v output 8.0v output 3.3v output 100 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 current (a) efficieny (%) efficiency v in = 12v 75 80 85 90 95 100 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 current (a) efficieny (%) efficiency v in = 24v 75 80 85 90 95 100 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 current (a) efficieny (%)
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 7 of 14 detailed description the AOZ1210 is a current-mode step down regulator with integrated high side nmos switch. it operates from a 4.5v to 27v input voltage range and supplies up to 2a of load current. the duty cycle can be adjusted from 6% to 85% allowing a wide range of output voltages. features include; enable control, power-on reset, input under v oltage lockout, ?ed internal soft-start and thermal shut down. the AOZ1210 is available in so-8 package. enable and soft start the AOZ1210 has an internal soft start feature to limit in-rush current and ensure the output voltage ramps up smoothly to the regulation voltage. a soft start process begins when the input voltage rises to 4.1v and voltage on en pin is high. in the soft start process, the output v oltage is typically ramped to regulation voltage in 4 ms. the 4.0 ms soft start time is set internally. if the enable function is not used, connect the en pin to v in . pulling en to ground will disable the AOZ1210. do not leave en open. the voltage on the en pin must be above 2.5 v to enable the AOZ1210. when voltage on en pin falls below 0.6v, the AOZ1210 is disabled. if an application circuit requires the AOZ1210 to be disabled, an open drain or open collector circuit should be used to interface with the en pin. steady-state operation under steady-state conditions, the converter operates in ?ed frequency and continuous-conduction mode (ccm). the AOZ1210 integrates an internal n-mosfet as the high-side switch. inductor current is sensed by amplifying the voltage drop across the drain to source of the high side power mosfet. since the n-mosfet requires a gate voltage higher than the input voltage, a boost capacitor connected between the lx and bst pins drives the gate. the boost capacitor is charged while lx is low. an internal 10 ? s witch from lx to gnd is used to ensure that lx is pulled to gnd even in the light load. output v oltage is divided down by the external voltage divider at the fb pin. the difference of the fb pin voltage and reference is ampli?d by the internal transconductance error ampli?r. the error voltage, which shows on the comp pin, is compared against the current signal. the current signal is the sum of inductor current signal and r amp compensation signal, at the pwm comparator input. if the current signal is less than the error voltage, the internal high-side switch is on. the inductor current ?ws from the input through the inductor to the output. when the current signal exceeds the error voltage, the high-side switch is off. the inductor current is freewheel- ing through the schottky diode to the output. switching frequency the AOZ1210 switching frequency is ?ed and set by an internal oscillator. the switching frequency is set to 370khz. output voltage programming output voltage can be set by feeding back the output to the fb pin with a resistor divider network. in the applica- tion circuit shown in figure 1. the resistor divider network includes r 1 and r 2 . t ypically, a design is started by picking a ?ed r 2 value and calculating the required r 1 value with equation below. some standard values for r 1 and r 2 for the most commonly used output voltages are listed in table 1. tab le 1. the combination of r 1 and r 2 should be large enough to av oid drawing excessive current from the output, which will cause power loss. protection features the AOZ1210 has multiple protection features to prevent system circuit damage under abnormal conditions. over current protection (ocp) the sensed inductor current signal is also used for over current protection. since the AOZ1210 employs peak current mode control, the comp pin voltage is propor- tional to the peak inductor current. the comp pin voltage is limited to be between 0.4v and 2.5v internally. the peak inductor current is automatically limited cycle by cycle. v o (v) r 1 (k ? ) r 2 (k ? ) 0.8 1.0 open 1.2 4.99 10 1.5 10 11.5 1.8 12.7 10.2 2.5 21.5 10 3.3 31.6 10 5.0 52.3 10 v o 0.8 1 r 1 r 2 ------- + ?? ?? ?? =
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 8 of 14 the cycle-by-cycle current limit threshold is internally set. when the load current reaches the current limit thresh- old, the cycle-by-cycle current limit circuit turns off the high side switch immediately to terminate the current duty cycle. the inductor current stops rising. the cycle- b y-cycle current limit protection directly limits inductor peak current. the average inductor current is also limited due to the limitation on the peak inductor current. when cycle-by-cycle current limit circuit is triggered, the output v oltage drops as the duty cycle decreases. the AOZ1210 has internal short circuit protection to protect itself from catastrophic failure under output short circuit conditions. the fb pin voltage is proportional to the output voltage. whenever the fb pin voltage is below 0.2v, the short circuit protection circuit is triggered. to prevent current limit running away when the comp pin v oltage is higher than 2.1v, the short circuit protection is also triggered. as a result, the converter is shut down and hiccups at a frequency equals to 1/16 of normal s witching frequency. the converter will start up via a soft start once the short circuit condition is resolved. in short circuit protection mode, the inductor average current is g reatly reduced because of the low hiccup frequency. po wer-on reset (por) a power-on reset circuit monitors the input voltage. when the input voltage exceeds 4.3v, the converter starts operation. when input voltage falls below 4.1v, the converter will stop switching. thermal protection an internal temperature sensor monitors the junction temperature. it shuts down the internal control circuit and high side nmos if the junction temperature exceeds 145?. the regulator will restart automatically under the control of soft-start circuit when the junction temperature decreases to 100?. application information the basic AOZ1210 application circuit is shown in figure 1. component selection is explained below. input capacitor the input capacitor (c 1 in figure 1) must be connected to the v in pin and gnd pin of the AOZ1210 to maintain steady input voltage and ?ter out the pulsing input current. the voltage rating of the input capacitor must be g reater than maximum input voltage + ripple voltage. the input ripple voltage can be approximated by equation below: since the input current is discontinuous in a buck converter, the current stress on the input capacitor is another concern when selecting the capacitor. for a buck circuit, the rms value of input capacitor current can be calculated by: if let m equal the conversion ratio: the relationship between the input capacitor rms current and voltage conversion ratio is calculated and shown in figure 2. it can be seen that when v o is half of v in , c in is under the worst current stress. the worst current stress on cin is 0.5 x i o . figure 2. i cin vs. voltage conversion ratio f or reliable operation and best performance, the input capacitors must have a current rating higher than i cin_rms at the worst operating conditions. ceramic capacitors are preferred for input capacitors because of their low esr and high ripple current rating. depending on the application circuits, other low esr tantalum capacitor or aluminum electrolytic capacitor may also be used. when selecting ceramic capacitors, x5r or x7r type dielectric ceramic capacitors are preferred for their better temperature and voltage characteristics. note that the ripple current rating from capacitor manufactures is based on certain amount of life time. further de-rating may be necessary for practical design requirement. inductor the inductor is used to supply constant current to output when it is driven by a switching voltage. for given input and output voltage, inductance and switching frequency together decide the inductor ripple current, which is, ? v in i o fc in ------------------ - 1 v o v in ---------- ?? ?? ?? v o v in ---------- = i cin_rms i o v o v in ---------- 1 v o v in ---------- ?? ?? ?? = v o v in ---------- m = 0 0.1 0.2 0.3 0.4 0.5 0 0.5 1 m i cin_rms (m) i o ? i l v o fl ----------- - 1 v o v in ---------- ?? ?? ?? =
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 9 of 14 the peak inductor current is: high inductance gives low inductor ripple current but requires larger size inductor to avoid saturation. low r ipple current reduces inductor core losses. it also reduces rms current through inductor and switches, which results in less conduction loss. when selecting the inductor, make sure it is able to handle the peak current without saturation even at the highest operating temperature. the inductor takes the highest current in a buck circuit. the conduction loss on inductor needs to be checked for thermal and ef?iency requirements. surface mount inductors in different shape and styles are av ailable from coilcraft, elytone and murata. shielded inductors are small and radiate less emi noise. but they cost more than unshielded inductors. the choice depends on emi requirement, price and size. output capacitor the output capacitor is selected based on the dc output v oltage rating, output ripple voltage speci?ation and r ipple current rating. the selected output capacitor must have a higher rated v oltage speci?ation than the maximum desired output v oltage including ripple. de-rating needs to be consid- ered for long term reliability. output ripple voltage speci?ation is another important f actor for selecting the output capacitor. in a buck con- ve r ter circuit, output ripple voltage is determined by inductor value, switching frequency, output capacitor v alue and esr. it can be calculated by the equation below: where; c o is output capacitor value and esr co is the equivalent series resistor of output capacitor. when low esr ceramic capacitor is used as output capacitor, the impedance of the capacitor at the switching frequency dominates. output ripple is mainly caused by capacitor value and inductor ripple current. the output r ipple voltage calculation can be simpli?d to: if the impedance of esr at switching frequency dominates, the output ripple voltage is mainly decided by capacitor esr and inductor ripple current. the output r ipple voltage calculation can be further simpli?d to: f or lower output ripple voltage across the entire operat- ing temperature range, x5r or x7r dielectric type of ceramic, or other low esr tantalum capacitor or aluminum electrolytic capacitor may also be used as out- put capacitors. in a buck converter, output capacitor current is continu- ous. the rms current of output capacitor is decided by the peak to peak inductor ripple current. it can be calcu- lated by: usually, the ripple current rating of the output capacitor is a smaller issue because of the low current stress. when the buck inductor is selected to be very small and induc- tor ripple current is high, output capacitor could be over- stressed. schottky diode selection the external freewheeling diode supplies the current to the inductor when the high side nmos switch is off. to reduce the losses due to the forward voltage drop and recovery of diode, a schottky diode is recommended. the maximum reverse voltage rating of the chosen schottky diode should be greater than the maximum input voltage, and the current rating should be greater than the maximum load current. loop compensation the AOZ1210 employs peak current mode control for easy use and fast transient response. peak current mode control eliminates the double pole effect of the output l&c ?ter. it greatly simpli?s the compensation loop design. with peak current mode control, the buck power stage can be simpli?d to be a one-pole and one-zero system in frequency domain. the pole is the dominant pole and can be calculated by: i lpeak i o ? i l 2 -------- - + = ? v o ? i l esr co 1 8 fc o -------------------------- - + ?? ?? = ? v o ? i l 1 8 fc o -------------------------- - ?? ?? = ? v o ? i l esr co = i co_rms ? i l 12 ---------- = f p 1 1 2 c o r l ----------------------------------- - =
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 10 of 14 the zero is a esr zero due to output capacitor and its esr. it is can be calculated by: where; c o is the output ?ter capacitor, r l is load resistor value, and esr co is the equivalent series resistance of output capacitor. the compensation design is actually to shape the converter close loop transfer function to get desired gain and phase. several different types of compensation network can be used for AOZ1210. for most cases, a series capacitor and resistor network connected to the comp pin sets the pole-zero and is adequate for a stable high-bandwidth control loop. in the AOZ1210, fb pin and comp pin are the inverting input and the output of internal transconductance error ampli?r. a series r and c compensation network connected to comp provides one pole and one zero. the pole is: where; g ea is the error ampli?r transconductance, which is 200 x 10 -6 a/v, g vea is the error ampli?r voltage the zero given by the external compensation network, capacitor c c (c5 in figure 1) and resistor r c (r1 in figure 1), is located at: to design the compensation circuit, a target crossover frequency f c f or close loop must be selected. the system crossover frequency is where the control loop has unity gain. the crossover frequency is also called the converter bandwidth. generally a higher bandwidth means faster response to load transient. however, the bandwidth should not be too high due to system stability concern. when designing the compensation loop, converter stability under all line and load condition must be considered. usually, it is recommended to set the bandwidth to be less than 1/10 of switching frequency. it is recommended to choose a crossover frequency less than 30khz. the strategy for choosing r c and c c is to set the cross ov er frequency with r c and set the compensator zero with c c . using selected crossover frequency, f c , to calculate r c : where; f c is desired crossover frequency, v fb is 0.8v, g ea is the error ampli?r transconductance, which is 200x10 -6 a/v, and g cs is the current sense circuit transconductance, which is 5.64 a/v the compensation capacitor c c and resistor r c together make a zero. this zero is put somewhere close to the dominate pole fp1 but lower than 1/5 of the selected crossover frequency. c c can is selected by: the equation above can also be simpli?d to: an easy-to-use application software which helps to design and simulate the compensation loop can be found at www .aosmd.com . thermal management and layout consideration in the AOZ1210 buck regulator circuit, high pulsing current ?ws through two circuit loops. the ?st loop starts from the input capacitors, to the v in pin, to the lx pins, to the ?ter inductor, to the output capacitor and load, and then returns to the input capacitor through g round. current ?ws in the ?st loop when the high side s witch is on. the second loop starts from inductor, to the output capacitors and load, to the gnd pin of the a oz1210, to the lx pins of the azo1210. current ?ws in the second loop when the low side diode is on. in pcb layout, minimizing the two loops area reduces the noise of this circuit and improves ef?iency. a ground plane is recommended to connect input capacitor, output capacitor, and gnd pin of the AOZ1210. in the AOZ1210 buck regulator circuit, the three major power dissipating components are the AOZ1210, e xternal diode and output inductor. the total power f z 1 1 2 c o esr co ------------------------------------------------- - = f p 2 g ea 2 c c g vea ------------------------------------------- = f z 2 1 2 c c r c ------------------------------------ - = f c 30 khz = r c f c v o v fb ----------- 2 c o g ea g cs ------------------------------ = c c 1.5 2 r c f p 1 ------------------------------------ = c c c o r l r c ---------------------- =
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 11 of 14 aoz 1210 /2 3 gnd 2 bst 1 lx 4 fb 6 en 7 vin 8 vbias 5 comp vo c2 c 2 r2 r1 rc cc vin l1 c4 cb c1 dissipation of converter circuit can be measured by input power minus output power. the power dissipation of inductor can be approximately calculated by output current and dcr of the inductor. the power dissipation of the diode is: the actual AOZ1210 junction temperature can be calculated with power dissipation in the AOZ1210 and thermal impedance from junction to ambient. the maximum junction temperature of AOZ1210 is 145?, which limits the maximum load current capability. the thermal performance of the AOZ1210 is strongly affected by the pcb layout. care should be taken by users during design process to ensure that the ic will operate under the recommended environmental conditions. several layout tips are listed below for the best electric and thermal performance. figure 3 is a layout example. 1. do not use thermal relief connection to the v in and the gnd pin. pour a maximized copper area to the gnd pin and the v in pin to help thermal dissipation. 2. input capacitor should be connected as close as possible to the v in and gnd pins. 3. make the current trace from lx pins to l to c o to gnd as short as possible. 4. pour copper plane on all unused board area and connect it to stable dc nodes, like v in , gnd or v out . 5. keep sensitive signal traces such as the trace connecting fb and comp pins away from the lx pins. p total _loss v in i in v o i o = p inductor _loss i o 2 r inductor 1.1 = p diode _loss i o v f 1 v o v in ---------- ?? ?? ?? = t junction p total _loss p inductor _loss () ja = t ambient ++ figure 3. layout example of the AOZ1210
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 12 of 14 pa ck ag e dimensions notes: 1. all dimensions are in millimeters. 2. dimensions are inclusive of plating 3. package body sizes exclude mold flash and gate burrs. mold flash at the non-lead sides should be less than 6 mils. 4. dimension l is measured in gauge plane. 5. controlling dimension is millimeter, converted inch dimensions are not necessarily exact. symbols a a1 a2 b c d e1 e e h l dimensions in millimeters min. 1.35 0.10 1.25 0.31 0.17 4.80 3.80 5.80 0.25 0.40 0 d c l h x 45 7 (4x) b 2.20 5.74 0.80 unit: mm 1.27 a1 a2 a 0.1 gauge plane seating plane 0.25 e 8 1 e1 e nom. 1.65 1.50 4.90 3.90 1.27 bsc 6.00 max. 1.75 0.25 1.65 0.51 0.25 5.00 4.00 6.20 0.50 1.27 8 symbols a a1 a2 b c d e1 e e h l dimensions in inches min. 0.053 0.004 0.049 0.012 0.007 0.189 0.150 0.228 0.010 0.016 0 nom. 0.065 0.059 0.193 0.154 0.050 bsc 0.236 max. 0.069 0.010 0.065 0.020 0.010 0.197 0.157 0.244 0.020 0.050 8
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 13 of 14 t ape and reel dimensions so-8 carrier tape so-8 reel so-8 tape leader/trailer & orientation tape size 12mm reel size ?30 m ?30.00 0.50 package so-8 (12mm) a0 6.40 0.10 b0 5.20 0.10 k0 2.10 0.10 d0 1.60 0.10 d1 1.50 0.10 e 12.00 0.10 e1 1.75 0.10 e2 5.50 0.10 p0 8.00 0.10 p1 4.00 0.10 p2 2.00 0.10 t 0.25 0.10 n ?7.00 0.10 k0 unit: mm b0 g m w1 s k h n w v r trailer tape 300mm min. or 75 empty pockets components tape orientation in pocket leader tape 500mm min. or 125 empty pockets a0 p1 p2 see note 5 see note 3 see note 3 feeding direction p0 e2 e1 e d0 t d1 w 13.00 0.30 w1 17.40 1.00 h ?3.00 +0.50/-0.20 k 10.60 s 2.00 0.50 g r v
a oz1210 rev. 1.3 december 2007 www.aosmd.com page 14 of 14 a oz1210 package marking z1 2 10ai fay part numbe r assembly lot code fab & assembly location year & week code wlt as used herein: 1. life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body or (b) support or sustain life, and (c) whose f ailure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a signi?ant injury of the user. 2. a critical component in any component of a life support, device, or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. this datasheet contains preliminary data; supplementary data may be published at a later date. alpha & omega semiconductor reserves the right to make changes at any time without notice. life support policy alpha & omega semiconductor products are not authorized for use as critical components in life support devices or systems.

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